Noise cancelling amplify-and-forward (in-band) relay with self-interference cancellation

ABSTRACT

The methods and systems for amplify-and-forward (in-band) relaying relate to beamforming techniques including receive and transmit beamforming for reducing self-interference, and improving Signal-to-Noise Ratio (SNR), or Signal to Interference plus Noise Ratio (SINR), of an incoming signal (to be relayed). The incoming signal is amplified and retransmitted simultaneously with the incoming signal, and over the same frequency band as that of an incoming signal.

CROSS-REFERENCE TO RELATED APPLICATIONS

The present application is related to and claims the benefit of theearliest available effective filing date(s) from the following listedapplication(s) (the “Related Applications”) (e.g., claims earliestavailable priority dates for other than provisional patent applicationsor claims benefits under 35 USC § 119(e) for provisional patentapplications, for any and all parent, grandparent, great-grandparent,etc. applications of the Related Application(s)). All subject matter ofthe Related Applications and of any and all parent, grandparent,great-grandparent, etc. applications of the Related Applications isincorporated herein by reference to the extent such subject matter isnot inconsistent herewith.

The present application is a continuation of U.S. patent applicationSer. No. 15/955,439 entitled “Noise Cancelling Amplify-and-Forward(In-Band) Relay With Self-Interference Cancellation,” filed Apr. 17,2018, which is hereby incorporated by reference herein in its entirety,which claims benefit under 35 USC § 119(e) for two provisionalapplications, U.S. Provisional Patent Application No. 62/487,274entitled “RF Beamforming”, filed Apr. 19, 2017 and U.S. ProvisionalPatent Application No. 62/487,273, entitled “Amplify and Forward Relaywith Self-Interference Cancellation”, filed Apr. 19, 2017, both of whichare incorporated herein by reference in their entirety.

BACKGROUND

Most wireless communication systems include a central node, such as acellular base-station, a WiFi access point, and/or an Internet of Thingsgateway communicating to a multitude of clients. In such configurations,it is desirable to increase the coverage area or range, and to removeblind spots. The straightforward approach is to increase transmit power.Simply increasing transmit power has several drawbacks. First, andforemost, transmission at high power levels increases the amount ofinterference to nearby nodes that may be reusing the same spectrum atthe same time. In addition, for resource-limited clients such as mobilephones, increasing the power level will have side effects such asbattery drainage and health implications for their users.

There is an ongoing need to enable transmitters to transmit at lowerpower levels.

SUMMARY

An apparatus and method includes embodiments directed to a noisecancelling amplify and forward relay. One embodiment is directed to anapparatus including a receive front-end including at least one receiveantenna operable at a first frequency band and responsive to an incomingradio frequency signal; an amplification stage coupled to the receivefront end, the amplification stage to amplify the incoming radiofrequency signal received at the receive front-end to provide anamplified incoming radio frequency signal; a transmit front-end coupledto the amplification stage to receive the amplified incoming radiofrequency signal, the transmit front-end including at least one transmitantenna operating at the first frequency band, the transmit front-end totransmit the amplified incoming radio frequency signal to a distantreceiver while the receive front-end receives the incoming radiofrequency signal; and a signal leakage filter stage coupled to thetransmit front-end, the signal leakage filter stage to reducetransmitted amplified incoming radio frequency signal leaked to thereceive front-end.

In one embodiment, the receive front-end includes a plurality of receiveantennas to perform receive beamforming, each of the plurality ofreceive antennas configured with a radio frequency beamforming tunablefilter.

In one embodiment, the plurality of receive antennas separately receivethe incoming radio frequency signal to enable radio frequency combiningand amplification to reduce signal leakage from the transmit front-endto the receive front-end via one or more tunable filters configured tocreate a wide-band receive null for the transmit front-end.

In one embodiment, the radio frequency beamforming tunable filters ineach of the plurality of receive antennas increase the beamforming gainto improve a signal-to-noise ration of the incoming radio frequencysignal.

In another embodiment, the transmit front-end includes a plurality oftransmit antennas to perform transmit beamforming, each of the pluralityof transmit antennas configured with a radio frequency beamformingfilter.

In one embodiment, the plurality of transmit antennas receive theamplified incoming radio frequency signal for transmit via a pluralityof filters that produce a transmit null over the receive front-end toreduce leakage from the transmit front-end to the receive front-end.

In one embodiment, the plurality of filters includes pairs of filters,wherein one of each pair of filters is being refreshed while another ofeach pair of filters is in use.

In one embodiment the plurality of transmit antennas receive theamplified incoming radio frequency signal for transmit via a pluralityof filters that focus the amplified incoming radio frequency signal fortransmit to a distant receiver.

In one embodiment, the apparatus includes a plurality of receiveantennas organized into two or more subsets of receive antennas toenable each receive antenna of the plurality of receive antennas to passthrough wide-band beamforming circuitry with a finite number of states.The finite number of states can include one or more of a 0° phase shift,a 180° phase shift and a disconnect.

In one embodiment, the amplified incoming radio frequency signal isformed as a composite signal combined from the output from the wide-bandbeamforming circuitry with the finite number of states.

In one embodiment, the composite signal includes a combined signalreceived via one or more output signals from each of the two or moresubsets of receive antennas, wherein each of the two or more subsets ofreceive antennas coupled to a frequency selective receive beamformingfilter.

The apparatus according to one embodiment includes a plurality oftransmit antennas that are grouped into two or more subsets of transmitantennas, each subset coupled to a frequency selective transmitbeamforming filter to produce a filtered signal, each frequencyselective beamforming filter coupled to divider and beamformingcircuitry with a finite number of states, wherein the beamformingcircuitry with the finite number of states is coupled to a transmitantenna to transmit a recombined amplified incoming radio frequencysignal.

In one embodiment, the apparatus also includes a signature signalgeneration circuit coupled to the transmit front-end, the signaturesignal generation circuit providing a signature signal included with theamplified incoming radio frequency signal for transmit, the signaturesignal to provide training for adapting to self-interference between thereceive front-end and the transmit front-end.

In one embodiment, the training for adapting to self-interferenceincludes one or more of blind channel estimation and analog echocancellation via a corrective signal inserted into the receivefront-end, the corrective signal created via a filtered transmit signal,the analog echo cancellation performed in at least one of RadioFrequency (RF), Intermediate Frequency (IF) and analog baseband.

In one embodiment, the apparatus transmit front-end and receive frontend include a plurality of transmit antennas and a plurality of receiveantennas, the plurality of transmit antennas symmetrically placed withrespect to the plurality of receive antennas to reduceself-interference.

In one embodiment, the plurality of receive antennas and the pluralityof transmit antennas each have two terminals for transmit and receiveover the same frequency band, the plurality of receive antennas and theplurality of transmit antennas being shared between transmit and receivefront-ends.

In one embodiment, the plurality of receive antennas and the pluralityof transmit antennas each have four terminals for transmit and receiveover at least two frequency bands, the plurality of receive antennas andthe plurality of transmit antennas being shared between transmit andreceive front-ends.

In one embodiment, the transmit front-end and the receive front endoperate with at least two frequency bands simultaneously relaying uplinkand downlink radio frequency signals in a Frequency Division Duplex(FDD) wireless network.

Another embodiment is directed to a method for relaying an incomingradio frequency signal including receiving the incoming radio frequencysignal at a receive front-end including at least one receive antennaoperable at a first frequency band and responsive to the incoming radiofrequency signal; amplifying the incoming radio frequency signal in anamplification stage coupled to the receive front end, the amplificationstage to amplify the incoming radio frequency signal received at thereceive front-end to provide an amplified incoming radio frequencysignal; transmitting the amplified incoming radio frequency signal via atransmit front-end coupled to the amplification stage, the transmitfront-end including at least one transmit antenna operating at the firstfrequency band, the transmit front-end transmitting the amplifiedincoming radio frequency signal to a distant receiver while the receivefront-end receives the incoming radio frequency signal; and reducingsignal leakage from the transmit front-end to the receive front-end viaproviding a self-interference cancellation channel, blind channelestimation and analog echo cancellation.

The foregoing summary is illustrative only and is not intended to be inany way limiting. In addition to the illustrative aspects, embodiments,and features described above, further aspects, embodiments, and featureswill become apparent by reference to the drawings and the followingdetailed description.

BRIEF DESCRIPTION OF THE FIGURES

FIG. 1 illustrates a transmitter and receiver in accordance with anembodiment.

FIG. 2 illustrates a frequency spectrum including superimposed signalsin accordance with an embodiment.

FIG. 3 illustrates transmitters and receivers without superimposedsignals in accordance with an embodiment.

FIG. 4 illustrates a relay structure for relaying FDD signals inaccordance with an embodiment.

FIG. 5 illustrates a receiver including a radio frequency balun and andtunable radio frequency attenuator for performing real multiplication inaccordance with an embodiment.

FIG. 6 illustrates a receiver that enables complex multiplication inaccordance with an embodiment.

FIG. 7 illustrates tunable delay element in accordance with anembodiment.

FIG. 8 illustrates a receiver implementation of a beamforming filteringin accordance with an embodiment.

FIG. 9 illustrates a receiver implementation of a RF beamforming filterin accordance with an embodiment.

FIG. 10 illustrates another receiver implementation of a RF beamformingfilter in accordance with an embodiment.

FIG. 11 illustrates another implementation of RF beamforming filter inaccordance with an embodiment.

FIG. 12 illustrates a transmitter implementation of an RF beamformingfilter in accordance with an embodiment.

FIG. 13 illustrates another transmitter implementation of an RFbeamforming filter in accordance with an embodiment.

FIG. 14 illustrates a hierarchical receiver implementation of an RFbeamforming filter in accordance with an embodiment.

FIG. 15 illustrates a symmetrical antenna structures with reducedcoupling between transmitter antenna and each of the two receiveantennas in accordance with an embodiment.

FIG. 16 illustrates a symmetrical antenna structure with reducedcoupling between transmitter antenna and each of two receive antennas inaccordance with an embodiment.

FIG. 17 illustrates an alternate version of a symmetrical antennastructure with reduced coupling between transmitter antenna and each oftwo receive antennas in accordance with an embodiment.

FIG. 18 illustrates antenna structures with frequency division duplex(FDD) relaying in which the antenna structures are implemented on theopposite sides of the relay box in accordance with an embodiment.

FIG. 19 illustrates an alternate version of antenna structures withfrequency division duplex (FDD) relaying in which the antenna structuresare implemented on the opposite sides of the relay box in accordancewith an embodiment.

FIG. 20 illustrates another alternate version of antenna structures withfrequency division duplex (FDD) relaying in which the antenna structuresare implemented on the opposite sides of the relay box in accordancewith an embodiment.

FIG. 21 illustrates a combined receiver and transmitter schematic ofarelay structure for relaying multiple-in-multiple-out (MIMO) signals inaccordance with an embodiment.

FIG. 22 illustrates a schematic of two auxiliary receivers that areshared in accordance with an embodiment.

FIG. 23 illustrates a schematic of an auxiliary receiver, denoted as“probing receiver” that is shared in accordance with an embodiment.

FIG. 24 illustrates a schematic of a training signal generator inaccordance with an embodiment.

FIG. 25 illustrates a block diagram showing a leakage effect in relationto blind estimation in accordance with an embodiment.

FIG. 26 illustrates another block diagram showing a leakage effect inrelation to blind estimation in accordance with an embodiment.

FIG. 27 illustrates a block diagram showing blind estimation inaccordance with an embodiment.

FIG. 28 illustrates a block diagram showing another version of blindestimation in accordance with an embodiment.

FIG. 29 illustrates a block diagram showing superimposed training andblind estimation in accordance with an embodiment.

FIG. 30 illustrates a block diagram of an RF beamforming filter inaccordance with an embodiment.

FIG. 31 illustrates a period sequence with low correlation propertiesfor the computation of the impulse response in accordance with anembodiment.

FIG. 32 illustrates a more detailed period sequence with low correlationproperties for the computation of the impulse response in accordancewith an embodiment.

FIG. 33 illustrates a block diagram of receivers adjusting the relativephase/magnitude (complex gain) of signals in accordance with anembodiment.

FIG. 34 illustrates a block diagram for adjusting phase at 0′ or 180′ ofsignals received from different receive antennas prior to RF combiningin accordance with an embodiment.

FIG. 35 illustrates another block diagram for adjusting phase at 0′ or180′ of signals received from different receive antennas prior to RFcombining in accordance with an embodiment.

FIG. 36 illustrates a graph of an LTE Demodulation Reference Signal(DRS) and a Sounding Reference Signal (SRS) as used in LTE appropriatefor embodiments.

FIGS. 37 and 38 illustrate embodiments of beamforming apparatus thatexchange information with legacy scheduler units in accordance withembodiments.

DETAILED DESCRIPTION

In the following detailed description, reference is made to theaccompanying drawings, which form a part hereof. In the drawings,similar symbols typically identify similar components, unless contextdictates otherwise. The illustrative embodiments described in thedetailed description, drawings, and claims are not meant to be limiting.Other embodiments may be utilized, and other changes may be made,without departing from the spirit or scope of the subject matterpresented here.

Embodiments herein address the need to enable transmitters to transmitat lower power levels by providing an external unit that amplifies andprovides a forward relay of their emitted signals.

Embodiments herein include amplify-and-forward relays that can be placedin locations in need of improved coverage. Thus, each relay receives theincoming signal from the central node, amplifies the incoming signal andemits the amplified signal to improve the coverage within aneighborhood. Embodiments herein provide amplify-and-forward relaystructures that: (1) provide a high gain (amplification) withoutoscillation, (2) improve the relayed signal in terms of its (end-to-end)Signal-to-Noise Ratio (SNR), or Signal to Interference plus Noise Ratio(SINR). In addition, the relaying operation includes embodiments that,including amplification and forwarding of the incoming signal, are fastenough such that the signal passing through the relay node appears asanother component of a multi-path propagation in the channel from thecentral node to a client (down-link), or vice versa, in the channel fromclient to the central node (uplink). As one of skill in the art willappreciate, a simple amplification of the incoming signal will alsoamplify the noise and amplify multi-user interference. Such noise andinterference embedded in the signal will degrade the performance,including throughput, and error rate, of the end-to-end link.

Embodiments described herein relate to an amplify and forward relaywhich operates as an interface between a central transmitter, such as acellular base-station, and client, such as mobile phones. The connectionbetween a central node and clients can be two-way, i.e., includingdownlink transmission from the central node to its clients and uplinktransmission from clients to the central node. Depending on theunderlying standard, the downlink and uplink connections are typicallymultiplexed either in the time domain or in the frequency domain.Amplify-and-forward relays according to embodiments herein handle bothdownlink and uplink connections.

Noise related to embodiments herein can be generated by feedback withinthe loop formed between transmit and receive front-ends of theamplify-and-forward relay, which can cause oscillation; and byamplification of noise embedded in the incoming signal, which canpotentially degrade the signal-to-noise ratio of the relayed signal. Thenoise can be a combination of thermal noise added by a first stage of areceive front-end, and multi-user interference caused by other nodesoperating over the same frequency band.

Embodiments described herein address noise generated and amplified andremain transparent to the operation of the signal that is being relayed.Accordingly, embodiments provide “receive beamforming”, “transmitbeamforming”, and “self-interference cancellation” by forming additionalpaths for the relayed signal with the property that a combination of thesignal flowing through these paths also cancel the self-interference,but do not cancel the incoming signal to be relayed.

Referring now to FIG. 1, a basic structure of transmitter and receiverunits is shown including a feedback structure in accordance with anembodiment is shown. Thus, a signature signal generated at baseband andmodulated to a radio frequency 100 is provided to an RF combiner 102 andamplified by amplifier 104. Each of the amplified signals is provided totransmitters 1-m, identified by transmitters, 106, 108 and 110.

Transmitters 1-m (106, 108, 110) also receive control signals 112 thatare also provided to receivers 1-m 114, 116, and 118. Each of receivers1, 2 through m include beamforming filters just like transmitters 1, 2through m. Outputs of receivers 114, 116 and 118 are provided to RFcombiner 120, which combines the signals and provides the output to RFdivider 122. RF divider 122 provides data to down-conversion andmeasurement of self-interference and measurement of SINR and generatingbeamforming control signals block 124. RF divider 122 also provides datato amplify block 126 and back to REF combiner 102.

Referring now to FIG. 2, a test signal is superimposed on an outgoingsignal and detected in a receiver chain for measuring self-interference.The results of the test signal superimposition are used to generatecontrol signals, such as control signals 112. As shown, carrierfrequency 210 is shown in the center of a spectrum of frequencies. Thespectrum of the incoming signal to be relayed, 212, the points ofobservation 214 and the spectrum of a superimposed test signal 216 aredisplayed. Thus, signature signals are superimposed within the frequencyband being relayed. The task of beamforming for nullingself-interference narrows down to reducing such signature signals at thereceiver front-end of the relay. Examples of such signature signalsinclude “low power OFDM signal with some tones left empty”, “chirpsignals”, “pseudo-random spreading codes”, orthogonal signals such as“Hadamard”, “Zadoff Chu sequence”, “Gold sequence”, and the like. Inmultiple-in-multiple-out (MIMO) operation, several signature signals arerequired which can be distinguishable from each other. The MIMOsignatures enable measuring the equivalent MIMO self-interferencechannel formed between transmit front-end and receive front-end of therelay node. The MIMO signatures can be generated using “timemultiplexing”, “frequency multiplexing”, or “code multiplexing such as,Zadoff Chu sequence of different parameters, or CDMA spreading codes ofa basic signature signal.

In addition to generating a signature signal, to cancel theself-interference, a receive structure evaluates the level of signaturesignal over an observation basis, such as a set of coordinates, which,collectively, capture the amount of self-interference over the frequencyband being relayed. Examples for such an observation coordinate systeminclude a discrete set of equally spaced points over the frequency band,or the impulse response of the self-interference channel in the timedomain.

Typically, an observation coordinate system and a signature generationcoordinate system are the same and are in the form of a set of equallyspaced points in the time and/or in the frequency domain. The “signaturegeneration coordinate system”, in conjunction with the “observationcoordinate system” enable measurement of the impulse response of theself-interference channel, such as the channel formed between transmitfront-end and receive front-end of a relay node.

Referring now to FIG. 3, an embodiment illustrates transmit and receivestructure without signature signals. A shown, a training signal for themeasurement of an impulse response of the self-interference channel issent in a separate time slot, time multiplexed. In all cases with orwithout using superimposed signature signals, the closed loop gain iscontrolled to avoid oscillation.

Blocks 306, 308, and Block 310, include transmitters 1-m with RFbeamforming filters. The structure includes control signal 312 which iscoupled to Block 324 which provides down conversion and measurement ofself interference and measurements of the desired signal Signal toInterference-plus-Noise Ratio (SINR) and generating beamforming controlsignals to maximize the desired SINR while avoiding loop isolation.Block 324 is also couple to RF divider 322 and amplify circuit 326.receiver circuits 314, 316, 318 represent receivers including RFbeamforming filter which received control signals from control signal312 RF combiner 320 is coupled to RF divider 322.

Another embodiment relies on separate (time multiplexed) trainingsignals to initialize the operation (measuring and cancelling ofself-interference), and then uses superimposed signature signals in thetracking phase. System can stop relaying and enter this initializationphase whenever the amount of self-interference is too high, oroscillation occurs in spite of closed loop gain control, or the gain toavoid oscillation is not enough for normal relaying operation to beeffective.

FIG. 4 shows a relay structure for relaying FDD signals, where F1 andF2, are uplink and downlink frequencies. Receive antennas are coupled toduplexers 420 and 426 and can receive over both bands. Transmitterantenna 408 sends over F1 and can receive at F2 (not shown), andtransmitter antenna 418 sends over F2 can receive over F1 (not shown).All received signals over F1 are combined after proper filtering,amplified and fed into transmit antenna operating over F1 408. Allreceived signals over F2 are combined after proper filtering andamplification and fed into transmit antenna operating over F2 418.Duplexer 420 separates F1 and F2 such that receiver 422 receives F1including beamforming filter at F1. Receiver 424 receives F2 includingare RF beamforming filter at F2. Receiver 422 is coupled to RF combinerat F1. Receiver 424 provides F2 to RF combiner 432. Receiver 428provides F1 to RF combiner 434 at F1. Receiver 430 provides F2 to RFcombiner 432.

RF dividers 436 and 438 receive frequencies F1 and F2, respectively andare both provided to block 440.

Block 440 provides down-conversion and measurement of self interferenceand measurement of the desired signal SINR and generates beamformingcontrol signals to maximize the desired SINR while avoiding looposcillation.

The signals received at RF dividers 436 and 438 are also provided toamplify circuits 442 and 444. The output of amplify circuits 442 and 444are provided to RF combiners 404 and 412.

Receive test band signals generated at baseband and modulated to the RFcarrier at the respective frequencies F1 and F2 are generated fromblocks 402 and 410, respectively and provided to RF combiners 404 and412. Next, the output of RF combiners 404 and 412 are provided toamplify circuits 406 and 416, which are then provided transmitter 1,408, and transmitter 2, 418.

Referring now to FIG. 5, a configuration for real multiplication(adjusting relative magnitude in different filter taps). As shown,receiver 502 is couple to entrance LNA 504, which is coupled totransformer 506, which is an RF Balun. Transformer 506 includes a switch508 for selecting one of the two ends of the secondary winding andprovide the output to tuneable RF attenuator 510. Thus, transformer 506is responsible for changing the sign and tunable attenuator 510 isresponsible for changing of magnitude.

FIG. 6 shows a configuration for complex multiplication (adjustingrelative magnitude/phase in different filter taps). Receiver 602 iscoupled to low noise amplifier (LNA) 604 which is coupled to quadraturehybrid coupler 606. Quadrature hybrid coupler 606 is responsible forgenerating two versions of the input signal (point 1) with a relativephase shift of 90° (points 3 and 4). Each transformer (an RF Balun) 608and 610 is responsible for changing the sign of its correspondingcomponent via switch 612 and switch 614, and each tunable attenuator 616and 618 is responsible for changing the magnitude of its correspondingcomponent and providing the output to RF combiner 620. The configurationof FIG. 6 is for complex multiplication and can be replaced withcommercially available complex multipliers such as a vector modulator,or concatenation of a tunable phase shifter and a tunable attenuator.

FIG. 7 illustrates a tunable delay element capable of generatingrelative delays from zero (very small) all the way to D(2^(n+1)−1) ininteger multiples of D. As shown, an RF input 702 is received by delays704, 706, 708 and 710, which are controlled with two position switches712 to provide RF output 714.

FIGS. 8, 9, 10, and 11 show some examples for the implementation of RFbeamforming filter at the receiver side.

Referring now to FIG. 8, receiver 1, 802 and receiver 2, 804 are coupledto low noise amplifiers (LNAs) 806 and 808, respectively. The output ofLNAs 806 and 808 are coupled to dividers 814 a-814 e and dividers 820a-820 e. Each of dividers 814 is separated by delays 826 a-826 d. Eachof dividers 820 is separated by delays 832 a-832 d. Each delay iscontrolled by a respective control 824 a-824 d and 822 a-822 d.

The outputs of dividers 814 a-814 e are provided to multipliers 812a-812 e. Likewise, the outputs of dividers 820 a-820 e are provided tomultipliers 828 a-828 e. Each of multipliers 812 a-812 e is controlledby control signals 810 a-810 e. Likewise, each of multipliers 828 a-828e is controlled by control signals 818 a-818 e. The outputs of each ofmultipliers 812 a-812 e and 828 a-828 e are provided to combiners 816a-816 e. The outputs of each of combiner 816 a-816 e are each providedto RF combiner 832, which provides output 840.

Referring now to FIG. 9, receiver 1 902 and receiver 2 904 are coupledto LNAs 906 and 908, respectively. The output of LNAs 906 and 908 arecoupled to dividers 914 a-914 e and dividers 920 a-920 e. Each ofdividers 914 is separated by delays 926 a-926 d. Each of dividers 920a-d is separated by delays 932 a-932 d. Unlike FIG. 8, there are nocontrol signals coupled to the dividers.

The outputs of dividers 914 a-914 e are provided to multipliers 912a-912 e. Likewise, the outputs of dividers 920 a-920 e are provided tomultipliers 928 a-928 e. Each of multipliers 912 a-912 e is controlledby control signals 910 a-910 e. Likewise, each of multipliers 928 a-928e is controlled by control signals 918 a-918 e. The outputs of each ofmultipliers 912 a-912 e and 928 a-928 e are provided to combiners 916a-916 e. The outputs of each of combiner 916 a-916 e are each providedto RF combiner 932, which provides output 940.

Referring now to FIG. 10, receiver 1, 1002 and receiver 2, 1004 arecoupled to LNAs 1006 and 1008, respectively. The output of LNAs 1006 and1008 are coupled to dividers 1014 a-1014 e and dividers 1020 a-1020 e.Each of dividers 1014 a-1014 e is separated by delays 1026 a-1026 d.Each of dividers 1020 a-1020 e is separated by delays 1032 a-1032 d.Unlike FIG. 8, there are no control signals coupled to the dividers.

The outputs of dividers 1014 a-1014 e are provided to multipliers 1012a-1012 e. Likewise, the outputs of dividers 1020 a-1020 e are providedto fixed attenuation and phase shift blocks 1028 a-1028 e. Each ofmultipliers 1012 a-1012 e is controlled by control signals 1010 a-1010e. The outputs of each of multipliers 1012 a-1012 e and fixedattenuation and phase shift blocks 1028 a-1028 e are provided tocombiners 1016 a-1016 e. The outputs of each combiner 1016 a-1016 e areeach provided to RF combiner 1032, which provides output 1040.

Referring now to FIG. 11, shown are LNAs 1106 and 1108 receiving signalsfrom receivers (not shown). LNA 1108 is coupled to complexmultiplication block 1110 and RF combiner 1120. LNA 1106 is showncoupled to dividers 1112 a-1112 e. Each of dividers 1112 a-1112 e isseparated by delays 1114 a-1114 d. Each divider 1112 a-1112 e is alsocoupled to a respective multiplication block that performs complex orreal or sign changes from 0°/180°.

Referring now to FIG. 12, an example of an implementation of an RFbeamforming filter from the transmitter side is illustrated. An RFtransmit signal 1202 is received at divider 1204 a. Dividers 1204 a-1204d are shown separated by delay blocks 1206 a-1206 d, respectively. Eachof delay blocks 1206 a-1206 d are provided with control signals 1010a-1010 d respectively. The outputs of dividers 1204 a-1204 d areprovided to respective multiplication blocks that perform complex orreal or sign changes from 0°/180° 1214 a-1214 d, and the output of delay1206 d is also provided to a multiplication block that performs complexor real or sign changes from 0°/180° 1214 e. Each of multiplicationblocks 1214 a-e is coupled to a control signal 1212 a-3. Each of theoutputs of multiplication block 1214 a-e are provided to RF combiner1220, which is coupled to amplifier 1230 and a transmit antenna 1240.

FIG. 13 illustrates another implementation of RF beamforming filter atthe transmitter side. An RF transmit signal 1302 is received at divider1304 a. Dividers 1304 a-1304 d are shown separated by delay blocks 1306a-1306 d, respectively. Each of delay blocks 1306 a-1306 d, unlike FIG.12, no control signals are coupled to the delays. The outputs ofdividers 1304 a-1304 d are provided to respective multiplication blocks1308 a-d that perform complex or real or sign changes from 0°/180° 1314a-1314 d, and the output of delay 1306 d is also provided to amultiplication block 1308 e that performs complex or real or signchanges from 0°/180° 1314 e. Each of multiplication blocks 1308 a-e iscoupled to a control signal 1310 a-e. Each of the outputs ofmultiplication block 1308 a-e are provided to RF combiner 1312, which iscoupled to amplifier 1314 and a transmit antenna 1320.

FIG. 14 illustrates an example for the implementation of RF beamformingfilter at the receiver side in a hierarchical structure. As shown,receivers 1-m identified by blocks 1410 and 1412 are coupled to receiveantennas 1402 and 1404 and identified as “Group 1”. Receivers m+1 and2m, identified by blocks 1414 and 1416 are coupled to receive antennas1406 and 1408 and identified as “Group 2”. Similar grouped receives arealso illustrated as group 3, group 4, group n and group n+1.

Receivers in Group 1 output to RF combiner 1418 and receivers in Group 2output to RF combiner 1420. RF combiner 1418 is coupled to amplifier1422, tunable filter 1426 and RF combiner 1430. Likewise RF combiner1420 is coupled to amplifier 1424, tunable filter 1428 and RF combiner1430. RF combiner 1430 is coupled to amplifier 1432, tunable filter 1434and RF combiner 1436, which provides an output.

Other grouped receivers such as group 3 and 4 and group n and n+1 canalso be added to the structure such that tunable filters 1428 a-1428 n,RF combiners 1430 a-1430 n, amplifiers 1432 a-1432 n and tunable filters1434 a-1434 n can each be output to an RF combiner as will beappreciated by one of skill in the art with the benefit of thisdisclosure.

FIG. 15 shows another example, for the implementation of RF beamformingfilter at the receiver side in a hierarchical structure. As shown, atransmitter 1 including RF beamforming filter 1502 and a transmitter 2including RF beamforming filter 1504 operate to relay signals receivedvia Receivers 1-4 1506, 1508, 1510 and 1512 separated into two groups,group 1 and group 2. Receivers 1506 and 1508 are coupled to RF combiner1514, and receivers 1510 and 1512 are coupled to RF combiner 1516. RFcombiner 1514 is coupled to amplifier 1518 and RF combiner 1516 iscoupled to amplifier 1520. The output of amplifiers 1518 and 1520 arerespectively coupled to tunable filters 1522 and 5244, and the outputsof the tunable filters is combined at RF combiner 1528. RF combiner 1528is coupled to amplifier 1530, which is coupled to RF divider 1532. RFdivider 1532 provides signals to down-conversion and measurement block1534 that provides down conversion and measurement of self-interferenceand measurement of the desired signal SINR and generates beamformingcontrol signals to maximize the desired SINR while avoiding looposcillation. The control signals 1542 generated by down-conversion andmeasurement block 1534 are provided to transmitters 1502 and 1504 fortransmission.

FIG. 16 illustrates an example of a symmetrical antenna structure withreduced coupling between transmitter antenna and each of the two receiveantennas. As shown, an X axis 1602 and a Y axis 1604 identify thelocation of receive antenna 1606 and 1608 with respect to transmitantenna 1610. FIG. 17 illustrates another example of a symmetricalantenna structure showing plane of symmetry 1700, receive antenna 1702and receive antenna 1704 and transmit antenna 1706 between the receiveantennas 1702 and 1704. The configurations shown in FIGS. 16 and 17 canbe extended horizontally (along the X axis) to include more antennas.FIG. 16 illustrates that the configuration can be also extendedvertically (along the Y axis) by stacking several layers of receiveantenna on either sides of a main layer, a main layer being the one thatincludes the transmit antenna in its center, and extends symmetricallyby placing receive antennas on the two sides of the transmit antennaalong the X axis. Receive antennas can be placed symmetrically aroundthe transmit antenna, 1610 or 1706, with transmit antenna at the centerof the array. The antenna at the center (transmit antenna) will have avery small coupling to the receive antenna placed on its left and rightsides along the main layer. In vertical extension, receive antennas inlayers above and below the main layer can be slightly rotated to reducethe coupling to the transmit antenna. Individual antennas in FIGS. 16and 17 can be implemented as patch structures.

Referring now to FIG. 18, an example for FDD relaying is illustrated,wherein antenna structures are implemented on the opposite sides of therelay box. Thus, front side 1802 and back side 1804 show symmetricantenna structures with receive antennas 1806 and 1808 placed on eitherside of transmit antenna 1810, and receive antennas 1812 and 1814 onother side of transmit antenna 1816. In FIG. 18, receive antennas 1812and 1814 receive at a first frequency, and receive antennas 1806 and1808 receive at a second frequency. Transmit antenna 1, 1810 transmitsat the first frequency, and transmit antenna 2, 1816 transmits at thesecond frequency.

FIG. 19 shows another embodiment for FDD relaying, wherein antennastructures are implemented on the opposite sides of the relay box. LikeFIG. 18, a front side 1902 holds a transmit/receive structure and a backside 1904 illustrates another transmit/receive structure. Front sidestructure includes receive antenna 1, 1906 receiving a first and asecond frequency, and receive antenna 2, 1908 also receiving at thefirst and second frequency, and transmit antenna 1 1910 transmitting atthe first frequency. Back side structure 1904 includes antenna 3, 1912receiving at the first and second frequencies and receive antenna 4,1914 also receiving at the first and second frequencies. Transmitantenna 2, 1916 is shown transmitting at the second frequency.

FIG. 20 shows another embodiment for FDD relaying, wherein antennastructures are implemented on the opposite sides of the relay box. Afront side 2002 holds a transmit/receive structure and a back side 2004illustrates another transmit/receive structure. Front side structureincludes receive antenna 1 2006 receiving a first and a secondfrequency, and receive antenna 2, 2008 also receiving at the first andsecond frequency, and transmit 1/receive 3 antenna 2010 transmitting atthe first frequency and receiving at the second frequency. Back sidestructure 2004 includes antenna 4, 2012 receiving at the first andsecond frequencies and receive antenna 5, 2014 also receiving at thefirst and second frequencies. Transmit antenna 2/Receive antenna 6 2016is shown transmitting at the second frequency, and receiving at thefirst frequency.

Referring now to FIG. 21 a generalization of the relay structure forrelaying MIMO signals is illustrated. As shown, signature signal type 1generated at baseband modulated to a first frequency 2102 is provided toRF combiner 2104 and followed by amplify 2106. Likewise, signaturesignal type 2 generated at baseband modulated to a second frequency 2108is provided to RF combiner 2110 and amplified at 2112.

The outputs of amplifier 2106 is provided to transmitter 1 operating afirst frequency 2124, and output of amplifier 2112 is provided totransmitter 2 operating at the first frequency 2126.

Likewise, signature signal type 3 operating at a second frequency 2114is provided to RF combiner 2117 and amplifier 2118, and signature signaltype 4 operating at the second frequency 2116 is provided to RF combiner2120 and amplifier 2122. Output of amplifier 2122 is provided totransmitter 2 operating at the second frequency 2130, and the output ofamplifier 2118 is provided to transmitter 1 operating at the secondfrequency 2128.

As shown there are eight receivers, identified as receiver 1, 2144,receiver 2, 2146, receiver 3, 2148, receiver 4, 2150, each of whichoperating at the first frequency; and receiver 5, 2132, receiver 6,2134, receiver 7, 2136, receiver 8, 2138, each of which operating at thesecond frequency. Receivers 1 and 2 are coupled to RF combiner 2152;receivers 3 and 4 are coupled to RF combiner 2154; receivers 5 and 6 arecoupled to RF combiner 2140; receivers 7 and 8 are coupled to RFcombiner 2142.

RF combiners 2152 and 2154 are coupled to RF dividers 2156 and 2158,respectively and the outputs of RF dividers 2156 and 2158 are providedto RF combiners 2104 and 2110 aw well as down-conversion and measurementblock 2160, which provides down-conversion and measurement ofself-interference and measurement of desired SINR and generatesbeamforming control signals. The outputs of RF combiner 2140 and 2142also are provided to down-conversion and measurement block 2160 as wellas to RF combiners 2120 and 2116, followed by amplifiers 2118 and 2122and to transmitter 1 operating at the second frequency 2128 andtransmitter 2 operating at the second frequency 2130.

FIG. 22 shows an embodiment in which two auxiliary receivers, denoted as“probing receiver 1, 2226” and “probing receiver 2, 2224” are shared forthe purpose of probing different received signals.

Specifically, four receivers including RF beamforming and tunablefilters 2202, 2204, 2206 and 2208 are shown coupled to RF divers 2210,2212, 2214, and 2216. The outputs of the RF dividers are provided to anRF switch 2218 and to RF combiner 2220. The output of RF combiner 2220is provided to 2222.

The output of RF divider is provided to transmitter(s) and to probingreceiver 2, 2224. RF switch 2218 is also provided to probing receiver 12226. The outputs of both probing receivers 1 and 2 are provided tocontrol unit 2228, which generates beamforming weights to maximize SINRwhile avoiding oscillation.

Probing receiver 1 2226 alternates among different receive antennas inorder to (sequentially in time) update the beamforming data relevant toeach of those antennas. Probing receiver 2 2224 measures the combinedsignal in parallel. This configuration enables maintaining synchronicityin relating the “cause of the change (adjustment in individual receivingfilters)” to their corresponding “effects (change in the combinedsignal)”.

FIG. 23 provides an embodiment in which only one auxiliary receiver,denoted as “probing receiver 2328” is shared for the purpose of probingdifferent received signals.

Specifically, four receivers including RF beamforming and tunablefilters 2302, 2304, 2306 and 2308 are shown coupled to RF divers 2310,2312, 2314, and 2316. The outputs of the RF dividers are provided to anRF switch 2318 and to RF combiner 3220. The output of RF combiner 2320is provided to RF switch 2326, which is also coupled to RF switch 2318.

The output of RF switch 2326 is provided to probing receiver 2328. Theoutput of probing receiver 2328 and then to control unit 2330, whichgenerates beamforming weights to maximize SINR while avoidingoscillation.

FIG. 24 shows the construction of a simple RF source for generating thetraining signal to be embedded in the transmit signal. Morespecifically, carrier 2402 is coupled to transformer 2404, which iscoupled to switch 2406 that is capable of selecting one of two ends oftransformer 2404 to enable 0 or 180 phase shift. The output of switch2406 is provided to bandpass filter centered at a carrier frequency 2408and the output provides a training signal 2410 such as a binary phaseshift keying (BPSK) with periodic repetition of a low correlationsequence, such as an Alexis sequence.

FIG. 25 illustrates a pictorial view of the leakage effect in relationto blind estimation setup. More specifically, a signal to be relayed2502 is coupled to filters D1, D2, D3 and D4, 2504, 2506, 2508 and 2510.Each of filters D1-4 are coupled to adders 2512, 2514, 2516 and 2518,followed by receiver 1, 2520, receiver 2, 2522, receiver 3, 2524, andreceiver 4, 2526. Each receiver is coupled to a tunable filter, 2528,2539, 2532 and 2534, respectively. Each tunable filter is coupled to RFcombiner 2536, which combines the outputs of each tunable filter. RFcombiner 2536 provides an output to transmitter 2538, which transmits tofilters 2540, 2542, 2544 and 2546, each of which provide additivesignals to adders 2512, 2514, 2516 and 2518 as described.

FIG. 26 shows another pictorial view of the leakage effect in relationto blind estimation setup. Specifically, signal to be relayed 2602 isprovided to filter D1 2604, which is provided to adder 2606, receiver2608 and tunable filter G1, 2610 and to RF combiner 2616. Transmitter2614 is shown receiving signals from RF combiner 2616 and transmittingto filter L1 2612. FIG. 26 also shows receivers 2, 2618, receiver 3,2626 and receiver 4, 2622, which are coupled to tunable filters 2624,2626 and 2628 and combined at RF combiner 2616.

FIG. 27 illustrates a blind estimation setup in accordance with anembodiment. As shown four receivers 2702, 2704, 2706 and 2708 arecoupled to tunable filters 2710, 2712, 2714 and 2716. Each of thetunable filters is coupled to an RF combiner 2720 and switch 2718. TheOutput of RF combiner 2720 is provided to transmitter 2730.

Switch 2718 is coupled to receiver 1, 2722, which providesdown-conversion and analog to digital conversion. Receiver 1 is coupledto processor 2724 for computing filters. Processor 2724 is also coupledto receiver 2 2726 which receives signals from transmitter 2730. Thus,two receivers 2722 and 2726 are used, one shared (2722) and onededicated (2726).

FIG. 28 shows another embodiment for a blind estimation setup. As shownfour receivers 2802, 2804, 2806 and 2808 are coupled to tunable filters2810, 2812, 2814 and 2816. Each of the tunable filters are coupled to anRF combiner 2820 and switch 2818. The output of RF combiner 2820 isprovided to transmitter 2830.

Switch 2818 is coupled to receiver 1, 2822, which providesdown-conversion and analog to digital conversion. Receiver 1 is coupledto processor 2824 for computing filters. Processor 2824 is also coupledto receiver 2, 2826 which receives signals from transmitter 2830. Thus,two receivers 2822 and 2826 are used, one shared (2822) and onededicated (2826).

FIG. 29 illustrates another embodiment combining the concepts of“superimposed training” and that of “blind estimation”. As shown fourreceivers 2902, 2904, 2906 and 2898 are coupled to tunable filters 2910,2912, 2914 and 2916. Each of the tunable filters are coupled to an RFcombiner 2920 and switch 2918. The output of RF combiner 2820 isprovided to transmitter 2930.

Switch 2818 is coupled to receiver 1, 2922, which providesdown-conversion and analog to digital conversion. Receiver 1 is coupledto processor 2924 for computing filters. Processor 2924 is also coupledto receiver 2, 2926 which receives signals from transmitter 2930. Thus,two receivers 2922 and 2926 are used, one shared (2922) and onededicated (2926). Unlike FIG. 28, FIG. 29 provides a simple source forgenerating a training signal 2940 coupled to a switch 2950 which can beprovided to transmitter 2930 for training.

Superimposed training signal can be activated as needed, for example,when the relay is idle, or be used as an auxiliary mechanism (inconjunction with the blind estimation technique) to facilitate the taskof estimation and compensation (tuning of filters).

FIG. 30 illustrates a method for the implementation of the RFbeamforming filter 3000 in accordance with an embodiment. As shown,input 3002 and signals g1-n, 3004, 3006, 3008, and 3010 are provided torespective tunable complex multipliers 1-n, 3018, 3020, 3022 and 3024,which can be vector modulators. Multipliers 1-n also receive signals viainput 3002 after a series of delays, 3012, 3014 and 3016 which can bebandpass SAW filters with a bandwidth equal or slightly higher than thatof the RF signal to be relayed. The output of the final delay 3016 isprovided as an output 3030.

FIG. 31 illustrates a periodic sequence 3100 with low correlationproperties for the computation of the impulse response. In particular,the provided example concerns the use of an Alexis sequence of length 323104 over time period T 3102. FIG. 31 shows the periodic repetition ofthe base sequence.

FIG. 32 illustrates another periodic sequence diagram 3200 with cyclicshifts 3202 over a computation window at time T 3204 and a periodicsequence 3210 with a computation window at time T+1 3206. Thus, FIG. 32shows two consecutive computational windows for extracting the impulseresponse.

Each computational window 3204 and 3206 show the base sequence, V1, andcyclic shifts of V1, denoted as V2, V3, . . . V32 3202. For simplicity,only 9 of such cyclic shifts are shown.

Sequence of output symbols (to be used for computing the impulseresponse) is denoted as [X1, X2, . . . X32, X33, . . . ] 3208.

Assuming the impulse response is the complex vector [I0, I1, I2, . . . ,I9, . . . , I32]. For simplicity, only the first nine components of theimpulse response are shown. The assumption is that the impulse responseis limited to 32 samples.

Inner product of vector Vi, i=1 to 32 (within the computation window attime T) with vector [X1, X2, . . . , X32] provide an estimate of thevalues of the impulse response [I0, I1, I2, . . . , I32].

Inner product of vector Vi, i=1 to 32 (within the computation window attime T+1) with vector [X2, X3, . . . , X33] provide an estimate of thevalues of the impulse response [I1, I2, . . . , I32, I0]. Theconstruction of subsequent computational windows will be appreciated bythose of skill in the art with the benefit of this disclosure.

Each computational window provides an estimate for the impulse response.To improve estimation accuracy, estimates obtained over consecutivecomputational windows are averaged.

In the provided example, computational windows (used for the detection)are shown to operate synchronized with the base sequence and its cyclicshifts used in transmission. In practice, the impulse response istypically composed of multiple zeros in its initial part. To improvecomputational efficiency, the effect of such zeros can be accounted forby cyclically delaying the computational windows with respect totransmitted base sequence. For example, if three of the initial valuesare known to be zero for certain, then the vectors used in thecomputational window at time T will be V3, V4, V5, . . . , V32, V1, V2.

For the provided example, if the length of the impulse response is knownto be less than 32, then, only a subset of vectors in the computationalwindows suffice for finding the impulse response. In the providedexample in FIG. 32, the inner product of V2 with [X1, X2, X3, . . . ,X32] required within the computational window at time T can be reused incomputing the inner product of V1 with [X2, X3, . . . , X32, X33]required within the computational window at time T+1. For thisparticular example, this is achieved by adding X1 and subtracting X33,i.e., if R(V1,T+1)=<V1, [X2, X3, . . . , X32, X33]> is the inner productin the window at time T+1, producing:R(V1,T+1)=+X1−X33+R(V2,T), similarly,R(V2,T+1)=+X1−X33+R(V3,T),R(V3,T+1)=+X1−X33+R(V4,T),R(V4,T+1)=+X1−X33+R(V5,T),R(V5,T+1)=+X1+X33+R(V6,T),R(V6,T+1)=−X1+X33+R(V7,T),

FIG. 33 shows an embodiment for adjusting the relative phase/magnitude(complex gain) of signals received from different receive antennas priorto RF combining. Thus, receivers 1-n, shown as 3302, 3304, and 3306 arecoupled to complex gain blocks 1-n, shown as 3308, 3310, and 3312. Theoutput of complex gain blocks are coupled to RF combiner 3314, which iscoupled to RF port of legacy transceiver 3316. In some embodiments,control signalling 3318 can be provided to RF combiner 3314.

FIG. 34 shows an embodiment for adjusting the phase at 0° or 180° or andrelative magnitude of signals received from different receive antennasprior to RF combining. As shown, receivers 3402 and 3406 are coupled RFtransformers 3408 and 3410, which are coupled to tunable RF attenuators3412 and 3414, which are coupled to RF combiner 3416. RF combiner 3416is coupled to RF port of legacy transceiver 3418. Control signalling3420 can be implemented to provide control from transceiver 3418 to RFcombiner 3416.

Referring to FIG. 35, another version of an embodiment for adjusting thephase at 0°/180° of signals received from different receive antennasprior to RF combining is provided. As shown, receivers 3502 and 3504 arecoupled to RF transformers 3506 and 3508, representing 2 or morereceivers and transformers. RF transformers 3506, 3508 are coupled to RFcombiner 3510, which is coupled to an RF port of a legacy transceiver3512. Again, possible control signals 3514 can be provided from thetransceiver to the RF combiner. An extreme case for adjusting relativemagnitude of a two state scenario of connect/disconnect can be realizedby implementing a three way switch (SP3T) at the secondary of the RFtransformers shown in FIG. 35, wherein upper and lower taps of the RFtransformer secondary provide the state of connect with 0°/180° relativephase shifts, and center tap of the RF transformer secondary (shown asgrounded) provide the disconnect state.

Referring now to FIG. 36, an embodiment relates to two beamformingstructures, alternating between training phase and utilization phase. Asshown, 1 subframe with two slots is shown on a time axis 3602 andfrequency axis 3604. Two users are shown, 3608 and 3610. The subframe isshown including data 3612, demodulation reference signal 3614 andsounding reference signal 3616.

The two beamforming structures can rely on the same set of antennas, butuse two different set of phase shifters (filters). In one embodiment,one set of phase shifters is being trained (connected to the auxiliaryreceiver), while the other set of phase shifters is kept fixed(connected to the main receiver). Switching between the two chains isperformed such that the operation of the receiver is not interrupted. Inone embodiment, switching is performed in early parts of the OFDM cyclicprefix.

Specifically, FIG. 36 shows the Demodulation Reference Signal (DRS) 3614and/or Sounding Reference Signal (SRS) 3616 used in LTE, which areexamples of preambles during training/tracking required in embodimentsdescribed herein. Another example is the preamble used in WiFi (802.11)for frequency mismatch estimation/correction.

Referring now to FIG. 37, embodiments are directed to beamformingapparatus that exchanges information with a legacy scheduler. As shown,even indexed phase 3740 includes receivers 3702, 3704 and 3706representing n receivers (1, 2 . . . n) are coupled to low noiseamplifiers and RF dividers 3708. RF dividers 3708 are coupled tobeamforming units 1 and 2. Beamforming unit 1 3710 can be connected to alegacy unit with or without control signalling 3714. Beamforming unit 23712 is shown under training coupled to an auxiliary receiverresponsible for training 3718. Beamforming unit 1 3710 is shown coupledto legacy receiver 3716 which provides signals to beamforming unity 3712and auxiliary receiver 3718. Odd indexed phase 3750 includes beamformingunit 1, 3720 connected to a legacy unit, beamforming unit 2 3722 undertraining, auxiliary receiver 3726 responsible for training and legacyreceiver 3728 that can provide possible control signalling 3730.

Referring now to FIG. 38, another version of embodiment directed tobeamforming apparatus that exchanges information with a legacy scheduleris illustrated. As shown, even indexed phase 3920 includes beamformingunits 1 and 2 3802 and 3804. Beamforming unit 1 3802 can be connected toa legacy unit with or without control signalling 3812. Beamforming unit2 3804 is shown under training coupled to an auxiliary receiverresponsible for training 3808. Beamforming unit 1 3802 is shown coupledto legacy receiver 3810 which provides signals to beamforming unit 13802, beforming in unit 2 3804 and to auxiliary receiver 3808. Bothbeamforming unit 2 3804 and auxiliary receiver 3808 responsible fortraining are coupled to legacy unit scheduling and channel stateinformation block 3806.

Odd indexed phase 3822 includes beamforming unit 1 3832 connected to alegacy unit, beamforming unit 2 3834 under training, auxiliary receiver3838 responsible for training and legacy receiver 3840 that can providepossible control signalling 3842. Legacy receiver 3840 can providesignals to auxiliary receiver 3838, and both beamforming units 3832 and3834. Beamforming unit 2 3834 and auxiliary receiver 3838 are showncoupled to legacy unit scheduling and channel state information block3836.

As described, FIGS. 37 and 38 show embodiments wherein the beamformingapparatus exchanges information with a legacy scheduler. Examples forsuch information exchanges include identifying which resource blocks areallocated to any particular uplink user and enabling beamformingapparatus to query the base-station to make adjustments in itsscheduling operation, and/or query the base-station to initiate soundingoperation.

Three main challenges hinder practical realization ofamplify-and-forward relays: (1) noise/interference amplification, (2)delay, and (3) cancellation of self-interference (to allow increasingthe relay gain without causing oscillation in the underlying closedloop).

To avoid the issue of noise/interference amplification, embodimentsdescribed herein rely on receive and/or transmit beamforming to firstimprove the quality of the signal to be relayed, and thereby compensatefor the subsequent amplification of noise/interference.

The requirement of low delay relaying is particularly challenging. Tomeet the delay requirement, all underlying (adaptive) filteringoperations (required in self-interference reduction) in some embodimentsare be performed in the analog domain. Embodiments herein are explainedin terms of filtering/amplification performed in the Radio Frequency(RF). Similar techniques can be also applied in the IntermediateFrequency (IF), and/or in analog base-band. In some embodiments,multiple stages of self-interference reduction modules are combined foroperating in RF, IF and analog base-band domains, as each of thesedomains has its own pros and cons in realizing the required adaptivefiltering.

Delay Requirements

Delay requirements, in the context of downlink channel and the case ofuplink channel can be similar. In the case of downlink, the central nodetransmits its outgoing signal, which will be received by both clientsand the amplify and forward relay unit. In some embodiments, an amplifyand forward relay amplifies its incoming signal and retransmits theamplified signal to provide coverage for clients in its neighborhood.Clients will receive the downlink signal through two paths, one path isdirectly from the central node, and the second path is through theamplify and forward relay. It is important that these two paths aresimultaneously received at the client side, otherwise, one path wouldact as interference to the other one. “Simultaneous” in this contextmeans within a relative delay that can be absorbed by the client nodemethod of channel equalization. In other words, the two paths shouldresemble paths formed in wireless transmission due to multi-pathpropagation.

Wireless standards have the capability built-in to deal with such areception of the same signal through multiple paths, and can jointlyequalize the signals received through such multiple paths. The delayspread among such a multitude of paths is the key factor that determinesif the receiving units will be able to jointly equalize these separatepaths, or not. For example, in standards based on Orthogonal FrequencyDivision Multiplex (OFDM), as long as the delay spread is less than theduration of OFDM cyclic prefix, the receiver sees the effects of suchpaths as a compound OFDM channel and can rely on the training signals tocompute the frequency response of the compound channel, and accordingly,can equalize the combination of multiple signal in a manner that theend-to-end link, over each OFDM tone, will be a single equalizedconstellation.

In the context of amplify-and-forward relay structures disclosed herein,this requirement entails that the delay incurred in the process ofrelaying should not exceed certain threshold. In contradistinction toknown methods of relaying, contradicts methods used in decode andforward relays which first recover the data by deploying a completereceive chain, and then retransmit the same data by deploying a completetransmit chain. In setups using complete transmit/receive chains, thebulk of the processing is performed in the digital domain, whichincludes down-conversion, equalization, demodulation and decoding (tasksof the receive chain), and then re-encoding, re-modulation,up-conversion and re-transmission (tasks of the transmit chain). Incontrast, in amplify and forward relay, the processing performed on thereceived signal should be in the analog domain to avoid excessive delay.

Processing tasks to accomplish embodiments herein include (1) activecancellation of the self-interference through filtering and constructionof an auxiliary signal which would be combined with the receive and/ortransmit analog signals to reduce the effect of the self-interference;and (2) receive and/or transmit beamforming for the purpose of reducingthe self-interference, improve the signal-to-noise ratio by directingthe respective transmit and/or receive antennas' beam(s) to focus on thecentral node, improve the signal-to-noise ratio by directing therespective transmit and/or receive antennas' beam(s) to focus on theclients, improve the signal-to-noise ratio by directing the respectivereceive antennas' beam(s) to avoid interference from neighboringtransmitters operating over the same spectrum.

To perform the above processing tasks, the embodiments described hereinrely on an auxiliary receive chain which is used to provide samples ofvarious incoming and outgoing signals for the purpose of monitoringrelevant signals, and accordingly informing a signal processing enginewhich in turn controls the filtering and beamforming tasks. Furthermore,to distinguish the self-interference signal from the incoming signal tobe relayed, some embodiments rely on embedding a known periodic sequencein the outgoing signal, which will be extracted by a correlator receiverand used to determine characteristics of the self-interference channel,i.e., channel from the transmit antenna(s) to receive antenna(s) withinthe same amplify and forward relay node.

To reduce the self-interference, in some embodiments, antennas aresymmetrically placed as illustrated in FIGS. 17-20. Each antenna has atleast one plane of symmetry. There are two types of such planes ofsymmetry. One type entails even symmetry, see, e.g., FIG. 17, whichmeans RF field components at points symmetrically located at the twosides of the plane of symmetry are positive mirrors of each other(mirror without sign change) with respect to the plane of symmetry.Another type entails odd symmetry, which means RF field components atpoints symmetrically located at the two sides of the plane of symmetryare negative mirrors of each other (mirror with a sign change) withrespect to the plane of symmetry. In some embodiments, a first set ofantennas have at least one plane of even symmetry, and a second set ofantennas have two orthogonal planes of symmetry, one with even symmetryand one with odd symmetry. The placement of antennas is such that theplane of even symmetry of the first set aligns (overlaps) with the planeof odd symmetry of the second set, and the resulting shared plane willbe orthogonal to the plane of even symmetry of the second set.

To reduce hardware complexity of beamforming, and at the same timesimplify the complexity of beamforming algorithm, some embodimentsherein rely on one-bit beam-formers, such as two phase values of 0° and180°. In some embodiments, such beamforming circuitry is cascaded withcircuitry for gain adjustment.

In one embodiment, corresponding to an extreme case of gain adjustmentcircuitry, the gain values are binary, i.e., the circuit is eitherconnected (passing its input signal to its output), or is disconnected(terminating the path such that no signal gets through). This bypassfeature is used to mimic the operation of maximum ratio combining inreceive beamforming, wherein incoming paths with low signal levels(which would degrade the overall Signal-to-Noise Ratio-SNR) arebypassed. Tasks of transmit and receive beamforming are performed toprovide a compromise among several factors, including boost in signalgain and SNR, nulling an external interfering signal, and creating anull for the self-interference path to improve isolation (reduceself-interference), thereby enabling an increase in the relay forwardgain without causing oscillation.

The simple beamforming structures explained above, i.e., with two states(0° phase shift, 180° phase shift) or with three states (0° phase shift,180° phase shift, bypass/disconnect), can be tuned using a sequentialsearch algorithm. For the purpose of the search algorithm, thebeamforming circuitries can be sequentially indexed by 1, 2, 3, . . . .In subsequent steps of the search algorithm, the states of beamformingcircuitries can be changed one-by-one (in the order of their indices),and in each case, the new state is kept if it results in an improvementin the underlying “figure of merit,” or it is reverted back, ifotherwise. “Figure of merit” is a vector with components representing:“receive beamforming gain (SNR)”,” “transmit beamforming gain”,“improvement in transmit/receive isolation (reduction inself-interference)” and “reduction in interference observed fromexternal nodes operating over the same spectrum, i.e., improvement inSignal-to-Noise-plus-Interference Ratio (SINR)”. The first priority isto improve transmit/receive isolation such that the relay can providethe desired level of forward gain, and once this condition is satisfied,other figures of merit mentioned above will be considered.

As mentioned earlier, in one embodiment, a signature signal will beembedded in the outgoing signal, which will be in turn extracted at thereceiving end of the relay to distinguish the self-interference channelfrom the channel carrying the incoming signal from the distanttransmitter (signal to be relayed). In an embodiment, the signaturesignal is selected as a sequence of ±1, which is repeated to create aperiodic sequence. An example is binary Alexis sequence. In someembodiments, the use of ±1 has been exploited in simplifying thehardware circuitry required for up-conversion of the signature signal.

Such circuitry relies on a transformer with a center-tapped secondary.See FIGS. 5 and 6, for example. The carrier is fed to the primary ofthis transformer, and the output (RF modulated signature signal) isextracted from the secondary of the transformer. Secondary is connectedto an RF switch, which selects one of the two outputs of the secondaryfollowing the sign of the signature signal. The resulting RF modulatedsignal is filtered to limit its spectrum occupancy to the ranges ofinterest in the relaying operation. In the case of an amplify andforward relay, transmitter generating/sending the signature signal, andthe receiver detecting the signature signal can be within the same unit.In an embodiment, this feature is exploited to implement a correlationreceiver for the signature signal at the RF front end of the receivingside as in FIG. 5 for example. In this embodiment, the correlationreceiver is in essence similar to the center tapped transformer used forup-conversion of the signature signal. In this case, if the transformerat the receiving side is switched with a delayed version of thesignature signal, with a delay that is equal to the delay of one of thepropagation paths in the self-interference channel, there will be a peakat the carrier at the output of the correlation receiver.

In an embodiment, by searching for successive delay values that resultin peaks in the output of the correlation receiver, and measuringmagnitude and phase of the resulting peaks, the system extracts theentire impulse response of the self-interference channel. This knowledgeis used to adjust the filter structures used in the cancellation ofself-interference.

In another embodiment, instead of separately extracting these components(corresponding to components of multi-path in the self-interferencechannel), the self-interference channel is measured throughpre-processing (this “pre-processing” is in essence pre-equalizationwith respect to the self-interference channel). In this case, at thetransmitter side, instead of transmitting one stream of the signaturesignal, multiple copies of the signature signals, after applying anappropriate relative delay and phase/magnitude adjustment to eachdelayed copy, are summed and the resulting waveform is RF modulated.See, for example, FIG. 21.

The relative delays and complex gain adjustments are set such that allthe multipath components align in time and add coherently at thereceiving side of the relay. In this case, the correlation receiver isoperated (“operated” means “switching of the receiver correlatortransformer”) with a fixed copy of the signature signal (“fixed copy”means a single copy of the signature signal with a given delay andmagnitude/phase), and the relative delays among components of thesignature signal forming the pre-processed signature signal at thetransmitting side, and their corresponding complex gain adjustments, areset with respect to the fixed signature signal switching the receivingtransformer. All parameters (relative delays and complex gain values)are adjusted to create one large peak at the output of the correlationreceiver, and adjustment are made to maximize the energy at this peak.The large peak is the result of aligning and coherently adding all themultipath components in the self-interference channel (throughpre-equalization). This strategy is used in some embodiments as analternative approach to compute the impulse response of theself-interference channel.

In some embodiments, the antennas used primarily for the purpose ofreception, transmit as well, and antennas used primarily for the purposeof transmission, receive as well. This allows an increase the effectivenumber of antennas involved in the beamforming operation.

Some of other key points behind various embodiments are as follows:

Insert low power training signals in the relayed signal to learn thecharacteristics of the self-interference channel (signal leaked fromtransmitter of the relay back to its receiver).

Use RF delay elements creating a delay equivalent (or close to) to onesample of Nyquist sampling frequency. For example, a 20 Mhz RF channelcorresponds to Nyquist base delay of 50 nsec.

Use multiple transmit antennas to relay the signal while forming atransmit null at the receive front-end of the relay.

Use multiple receive antennas to enable receive beamforming for thepurposes of: Reducing self-interference through receive beamforming (tonull the self-interference channel), i.e., create a receive null withrespect to the relay transmit front-end, and improve the reception(signal to noise ratio) of the (desired) incoming signal prior toamplifying/relaying it.

One embodiment is for relaying Frequency Division Duplex (FDD) signals.FDD systems use two distinct frequency bands to send and to receive. Anembodiment relies on six antennas to relay a (SISO) FDD signal, such asthose shown in FIGS. 18, 19 and 20. These antennas are divided intogroups:

Set A1: Two antennas are mainly responsible for receiving over the firstfrequency band, F1. In an enhanced version, these two antennas canreceive over F2, or transmit over F2.

Set A2: Two antennas are mainly responsible for receiving over thesecond frequency band, F2. In an enhanced version, these two antennascan receive over F1, or transmit over F1.

B1: One antenna is mainly responsible for transmitting over the firstfrequency band, F1. In an enhanced version, this antenna can receiveover F2.

B2: One antenna is mainly responsible for transmitting over the secondfrequency band, F2. In an enhanced version, this antenna can receiveover F1.

In one embodiment, sets A1 and A2 act as receiver over F1 and F2; andsets B1 and B2 act as a transmitter over F1 and F2, respectively.

In another embodiment, sets A1 and A2 are comprised of many moreantennas, for example 100 antennas, each equipped with a simplebeamforming circuitry which, e.g., applies a one-bit phase adjustment,i.e., (0°,180°) phase adjustment, to the signal received by itsrespective antenna prior to signal combining. In a slightly moreadvanced version, the beamforming circuitry can be associated with eachantenna is capable of (0°, 180°) phase adjustment in a “connectedstate”, or terminating the signal in a “disconnected state”. Antennasthat are in the disconnected states will not contribute to the combinedsignal. The disconnecting the signal of any particular antenna fromentering the combiner allows to improve the effective signal-to-noiseratio by mimicking the operation used in maximum ratio combining. Thismeans, the signals of antennas that have a low signal-to-noise ratiowill be simply dropped prior to combining.

As will be appreciated by those of skill in the art, various otherconfigurations, including a large number of antennas in sets A1, A2, B1and B2, equipped with a combination of simple circuitries explainedabove for beamforming, and more complex beamforming circuitries foradjustment of phase and/or magnitude will be possible. In one embodimentappropriate for installation in locations that would allow large formfactors, sets A1 and A2 include hundreds of antennas and sets B1 and B2are composed of a smaller number of antennas (e.g., less than 10). Theuse of a large number of antennas allows satisfying the objectives of:“beamforming for the purpose of nulling self-interference,” as well as“realizing advantages of traditional beamforming for improving signalgain and/or reducing multi-user interference”.

Embodiments herein can be generalized to Multiple-Input Multiple-Output(MIMO) antenna structures based on the following principles: (1) In caseof MIMO, the basic structure is repeated M times, M is the number ofantennas in the M×M MIMO; (2) M transmit antennas are equipped with Mdistinguishable signature signals, such as sinusoidal separated in timeand/or in frequency, and (3) received signals are combined and relayedsuch that all signature signals are nulled.

Another embodiment disclosed herein relies on Blind Channel Estimationfor training (initial nulling of self-interference) and/or tracking(gradual readjustments upon completion of the training phase). Trackingincludes following time variations in the self-interference channel,and/or time variations in the channels related to the primary signals(used for the purpose of traditional beamforming towards improvingsignal gain and/or reducing multi-user interference). In embodimentsrelying on Blind Channel Estimation, instead of embedding a trainingsignal in the signal to be relayed, the task of channel estimation isperformed blindly by relying on some auxiliary receivers, called“probing receiver(s)”, and using the correspondence between transmit andreceive signals (in baseband) to estimate the impulse response of theleakage paths, and accordingly adjust the beamforming filters (see,e.g., FIGS. 25-29).

Wireless systems, such as LTE, benefit from beamforming for the purposeof maximizing the signal strength, minimizing interference, or acombination of the two, e.g., maximizing theSignal-to-Interference-Plus-Noise Ratio (SINR). Similar benefits can berealized for relayed signals. In addition to amplify-and-forward relays,embodiments for beamforming at the RF level can be integrated withlegacy setups. This would be an add-on component operating transparentto the operation of the legacy transceiver, while improving itsperformance. In the following, these techniques are explained in thecontext of interface to legacy setups, but similar setups apply to thecase of beamforming at the RF front-end for an amplify-and-forwardrelay. If a legacy system has multiple antennas, each of such legacyantennas can be enhanced by the addition of RF beamforming. Beamformingcan be learned in the uplink, which can be applied to uplink (FDDsetups), or to both uplink and downlink (TDD setups) relying on channelreciprocity. In the latter case, the downlink beamforming is adjustedbased on storing the selected beam for each particular client, which arethen selected knowing the client(s) targeted in a downlink transmissionphase.

Receive Beamforming

Hereafter, embodiments in the context of receive beamforming in theuplink.

Receive beamforming is achieved by one or more, or the combination of(1) adjusting the relative phase/magnitude (complex gain) of signalsreceived from different receive antennas prior to RF combining, such asFIG. 33; (2) adjusting the phase at 0° or 180° and relative magnitude ofsignals received from different receive antennas prior to RF combining,such as shown in FIG. 34. Note that the relative magnitude, in itsextreme case of one-bit resolution, means the signal is either connectedor is disconnected.

Adjusting the phase at 0° or 180° of signals received from differentreceive antennas prior to RF combining is described above with respectto FIG. 35.

Transparent Training/Tracking

Embodiments herein relate to perform training for the computation ofbeamforming strategy such that the operation of the legacy receiver isnot affected. This operation can be performed by training/trackingduring receiver idle time (when there are no signals in the air).

Training/tracking is performed during preambles that are not used forchannel training/equalization. An example is the preamble used in WiFi(802.11) for frequency mismatch estimation/correction, or theDemodulation Reference Signal (DRS) and/or Sounding Reference Signal(SRS) used in LTE. See, for example, FIG. 36.

Training/tracking can be performed during cyclic prefix of the OFDM orSC-FDMA structure (in parts that will be discarded by the legacytransceiver).

Embodiments herein include two beamforming structures, alternatingbetween training phase and utilization phase. The two beamformingstructures can rely on the same set of antennas, but use two differentset of phase shifters (filters). In this embodiment, one set of phaseshifters is being trained (connected to the auxiliary receiver), whilethe other set of phase shifters is kept fixed (connected to the mainreceiver). The “switched training” is described with respect to FIGS. 37and 38. Switching between the two chains is performed such that theoperation of the receiver is not interrupted. In one embodiment,switching is performed in early parts of the cyclic prefix.

Using Legacy Preambles for Training

In order to avoid modification to legacy standard, embodiments hereinrely on preamble structures used in legacy system, in particularpreamble used for channel estimation, to compute the beamformingstrategy. In particular, in standards based on OFDM and its variants(such as SC-FDMA used in LTE Uplink), training signals are transmittedthat have equal magnitude in all their occupied tones. These signals aretypically used for the purpose of channel estimation. Upon compensatingfor the phase in successive tones (frequency segments), the sequence ofthe equalized tones will be separated into two parts: (1) Slowlychanging part specifies the signal gain (channel magnitude); and (2)Fast changing part specifies the noise plus interference.

These two parts can be separated by a simple filtering of the successivevalues of the equalized tones.

In case of switched training, the preamble as well as data part ofsuccessive OFDM (or SC-FDMA) symbols can be used.

In a preferred embodiment for LTE uplink, DRS and/or SRS, possibly inaddition to in-between SC-FDMA data symbols are used. In DRS (as well asin successive SC-FDMA symbols within a “slot”), regardless of userscheduling, it is known that at least 12 successive carriers arereceived from the same user, and, as a result, averaging over segmentsof length 12 can be used to estimate the signal and the interference.Then, a “min-max” (or an “averaging”) criterion can be used to selectthe best antenna pattern.

In LTE uplink, there are two reference signals as shown in FIG. 36. Thedemodulation reference signal can be separated into consecutive segmentsof tones corresponding to different transmitters sending in the uplinkusing SC-FDMA. Then, the signal power (channel gain) as well as theinterference level for each of these consecutive segments is computed(for different antenna patterns) and the best pattern is selected. Thebest pattern can be selected to balance the quality of the signalsreceived from different clients, i.e., maximize the minimum SINR (inorder to accommodate transmitters at the cell edge, or those in fading).

In one embodiment, switched training (as shown in FIGS. 37 and 38) inconjunction with 0°/180° phase shift structure as shown in FIGS. 33 and34 is used. During the cyclic prefix of the reference signals, or duringthe data part (see FIG. 38), the antennas' phases are sequentiallyswitched by 180°, the outcome of switching is computed over thesubsequent reference signal, and the applied phase shift isaccepted/rejected depending on the observed outcome. The latest bestpattern for each uplink transmitter is stored and used to initialize thesearch next time the same client is scheduled.

In one embodiment, the beamforming apparatus listens and interprets someof the control signalling exchanged between its associated base-stationand legacy units being served by this base-station. In particular, thisallows the beamforming apparatus to act synchronous with the timing ofthe legacy unit, for example, to extract the start of the demodulationreference signal, and accordingly adjust its switching and beamselection strategy.

In another embodiment as shown in FIG. 38, the beamforming apparatusexchanges information with the legacy scheduler. Examples for suchinformation exchanges include (1) knowing which resource blocks areallocated to any particular client in the uplink; and (2) beamformingapparatus can ask the base-station to make adjustments in its schedulingoperation, and/or ask the base-station to initiate the soundingoperation.

While various aspects and embodiments have been disclosed herein, otheraspects and embodiments will be apparent to those skilled in the art.The various aspects and embodiments disclosed herein are for purposes ofillustration and are not intended to be limiting, with the true scopeand spirit being indicated by the following claims.

The invention claimed is:
 1. An apparatus comprising: a receivefront-end including a plurality of receive antennas configured toperform receive beamforming, wherein the plurality of receive antennasinclude a respective first plurality of radio frequency beamformingfilters, and wherein the receive front-end operates within at least afirst frequency band and is responsive to an incoming radio frequencysignal; an amplification stage coupled to the receive front-end, theamplification stage configured to amplify the incoming radio frequencysignal received at the receive front-end and to produce an amplifiedincoming radio frequency signal; a transmit front-end coupled to theamplification stage to receive the amplified incoming radio frequencysignal, wherein the transmit front-end includes a plurality of transmitantennas configured to perform transmit beamforming, wherein theplurality of transmit antennas include a respective second plurality ofradio frequency beamforming filters, and wherein the transmit front-endoperates within at least the first frequency band, the transmitfront-end being configured to transmit the amplified incoming radiofrequency signal to a distant receiver while the receive front-end isreceiving the incoming radio frequency signal; an analog echocancellation path, wherein a corrective signal is inserted into thereceive front-end, wherein the corrective signal is created via afiltered transmit signal, and wherein analog echo cancellation isperformed in at least one of a radio frequency domain, an intermediatefrequency domain, and an analog baseband domain; a signature signalgeneration circuit coupled to the transmit front-end, wherein thesignature signal generation circuit is configured to generate asignature signal that is included within the first frequency band inwhich the amplified incoming radio frequency signal is to betransmitted, and wherein the signature signal includes a periodicsequence to distinguish the signature signal from the transmittedamplified incoming radio frequency signal; and a self-interferencemeasurement circuit configured to generate beamforming control signalsto control at least one of (i) analog echo cancellation, (ii) theplurality of receive antennas including the respective first pluralityof radio frequency beamforming filters or (iii) the plurality oftransmit antennas including the respective second plurality of radiofrequency beamforming filters, wherein the self-interference measurementcircuit is further configured to (i) detect the signature signal presentat the receive front-end to measure self-interference and (ii) to adaptan echo cancellation filter; and generate the beamforming controlsignals based on the detected signature signal, such that an amount ofthe transmitted amplified incoming radio frequency signal and thesignature signal leaked from the transmit front-end back to the receivefront-end is reduced.
 2. The apparatus of claim 1, wherein each of theplurality of receive antennas separately receives the incoming radiofrequency signal to enable radio frequency combining and amplification,and wherein the respective first plurality of radio frequencybeamforming filters is configured to create a receive null with respectto the transmit front-end so as to reduce a signal leakage from thetransmit front-end back to the receive front-end; and wherein theplurality of transmit antennas receives the amplified incoming radiofrequency signal for transmit to the distant receiver via the respectivesecond plurality of radio frequency beamforming filters that areconfigured to produce a transmit null with respect to the receivefront-end so as to reduce a signal leakage from the transmit front-endback to the receive front-end.
 3. The apparatus of claim 1, wherein eachof the respective first plurality of radio frequency beamforming filtersin the receive front-end is configured to increase a beamforming gain toimprove a signal-to-noise ratio of the amplified incoming radiofrequency signal.
 4. The apparatus of claim 1, wherein each of therespective first plurality of radio frequency beamforming filters in thereceive front-end or each of the respective second plurality of radiofrequency beamforming filters in the transmit front-end has at least twostates, wherein the at least two states include a 0° phase shift and a180° phase shift, and wherein each filter having the at least two statesis configured such that the at least two states are controlledselectively to improve isolation between the receive front-end and thetransmit front-end.
 5. The apparatus of claim 1, further comprising atraining signal generation circuit coupled to the transmit front-end,wherein the training signal generation circuit is configured to generatea training signal that is included with the amplified incoming radiofrequency signal for transmit, and wherein the training signal providesinitial training for adapting to self-interference between the receivefront-end and the transmit front-end.
 6. The apparatus of claim 1,wherein the transmit front-end and the receive front-end operate withinat least two frequency bands simultaneously relaying uplink and downlinkradio frequency signals in a Frequency Division Duplex (FDD) wirelessnetwork.
 7. The apparatus of claim 1, wherein each of the the pluralityof receive antennas includes a corresponding radio frequency beamformingfilter that has a connected state and a disconnected state.
 8. Theapparatus of claim 7, wherein one or more of the radio frequencybeamforming filters in the at least two sets of receive antennas are inthe disconnected state to prevent signals from corresponding one or morereceive antennas from entering a radio frequency combiner to improvesignal-to-noise ratio.
 9. The apparatus of claim 1, wherein theself-interference measurement circuit is further configured to evaluatea level of the signature signal over an observation basis to measure anamount of self-interference over the first frequency band.
 10. Theapparatus of claim 1, wherein the signature signal includes a periodicsequence to distinguish the signature signal from the transmittedamplified incoming radio frequency signal, wherein the periodic sequencecomprises a binary Alexis sequence, and wherein the self-interferencemeasurement circuit is further configured to evaluate a level of thesignature signal over a given observation basis to measure an amount ofself-interference over the first frequency band.
 11. The apparatus ofclaim 1, wherein the signature signal is multiplexed in time domain withthe wireless signal to be relayed.
 12. The apparatus of claim 1, whereintraining for at least one component contributing to self-interferencereduction is performed using blind training and/or blind trackingtechniques.
 13. The apparatus of claim 1, wherein training for at leastone component contributing to self-interference reduction relies ontraining signals included, as part of the underlying industry standard,in the structure of wireless signal to be relayed.
 14. The apparatus ofclaim 1, wherein at least one of a plurality of filters contributing toself-interference reduction includes pairs of filters, wherein one ofeach pair of filters is refreshed while another of each pair of filtersis in use.
 15. The apparatus of claim 14, wherein switching betweenfilter pairs is performed during a cyclic prefix of an OrthogonalFrequency-Division Multiplexing signal to be relayed.
 16. The apparatusof claim 1, wherein, the transmit front-end and receive front-endinclude a plurality of transmit antennas and a plurality of receiveantennas, the plurality of transmit antennas symmetrically placed withrespect to the plurality of receive antennas to reduceself-interference.
 17. The apparatus of claim 1, wherein the pluralityof receive antennas and the plurality of transmit antennas each have twoterminals for transmit and receive over a same frequency band, theplurality of receive antennas and the plurality of transmit antennasbeing shared between the transmit front-end and the receive front-end.18. The apparatus of claim 1, the plurality of receive antennas and theplurality of transmit antennas each have four terminals for transmit andreceive over at least two frequency bands, the plurality of receiveantennas and the plurality of transmit antennas being shared between thetransmit front-end and the receive front-end.
 19. The apparatus of claim1, wherein blind channel estimation is utilized for at least one oftraining, including initial nulling of self-interference, and tracking,including gradual readjustments upon completion of a training phase. 20.The apparatus of claim 1, wherein channel estimation is performed by aplurality of auxiliary probing receivers to estimate an impulse responseof leakage paths and accordingly adjust the radio frequency beamformingfilters and the echo cancellation filter.